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Linear Motor and Linear Stepper Motors

What is a Linear Motor?

Author: Alexis Gibrault

A linear motor is-simply speaking-an electric motor that uses a linear force mechanism to generate the power needed for a said application. In contrast to a rotational electric motor (found in automobiles, appliances, and commonly-used electrical equipment), a linear motor generates its energy output through exclusively linear scientific principles; i.e. there is no torque or rotation to produce accelerated force through the electrical current magnetic field relationship. Linear motors are used for a variety of purposes, which include high velocity trains, military weaponry, spacecraft exploration, robotic technologies, medical advancement, and automated engineering systems whose job is to produce mass amounts of a specified product.

There are two basic types of linear motors: low-acceleration and high-acceleration. Low-acceleration motors are typically used for applications in which endurance is favored over high bursts of electromechanical power or energy. These types of linear motors are engineered for Maglev trains, automated applications systems, etc. High-acceleration motors are the more common of the two, and produce higher velocity outputs for shorter amounts of time; such as used in firearms, military equipment, spacecraft propulsion, and the like. Low-acceleration linear motors are designed to accelerate an object up to a continuous stabile speed, while high-acceleration linear motors will accelerate an object up to a very high speed and then release the object. Typically, the low-acceleration linear motor will be engineered with one winding system on one side of the motor and magnets on the other side to create the electromagnetic repulsion necessary for successful application force; this is called linear synchronous design. The high-acceleration linear motor will generally be constructed of a three-phase winding on one side and a conductor plate on the other side of the motor to meet the intended engineering objective; this is called linear induction design.

Linear motors offer a number of advantages in this ever-evolving technological world. Whether the high power application your company or organization requires necessitates a low- or high-accelerated lateral motor system, linear motors assure faster acceleration and higher velocities as well as higher success rates in automated accuracy, repeatability, and long-term reliability.

About the Author:

Alexis Gibrault has written a number of informative articles on linear motors, types, and uses, as well as discussions on other facets of technology engineering. For more information on linear motors and examples of, please visit: Airex Corporation Linear Motors

Article Source: http://www.articlesbase.com/technology-articles/what-is-a-linear-motor-65122.html

Linear Stepper Motors Technology

Author: Gordon Petten

A linear induction motor is made up of an inductor which is made of individual cores with a concentrated polyphase. Linear motors can be directly substituted for ball screw drives, hydraulic drives, pneumatic drives, or cam drives.

A linear induction motor is basically what is referred to by experts as a "rotating squirrel cage" induction motor. The difference is that the motor is opened out flat. Instead of producing rotary torque from a cylindrical machine it produces linear force from a flat machine. The shape and the way it produces motion is changed, however it is still the same as its cylindrical counterpart. There are no moving parts, however and most experts don't like that. It does have a silent operation and reduced maintenance as well as a compact size, which appeals many engineers. There is also a universal agreement that it has an ease of control and installation. These are all important considerations when thinking about what type of device you want to create. The linear induction motor thrusts ratio varies depending mainly on the size and rating. Speeds of the linear induction motor vary from zero to many meters per second. Speed can be controlled. Stopping, starting and reversing are all easy. Linear induction motors are improving constantly and with improved control, lower life cycle cost, reduced maintenance and higher performance they are becoming the choice of the experts. Linear motors are simple to control and easy to use. They have a fast response and high acceleration. Their speed is not dependant on contact friction so it is easier to pick up speed quickly.

Stepper motors are a special kind of motor that moves in discrete steps. When one set of windings is energized the motor moves a step in one direction and when another set of windings is energized the motor moves a step in the other direction. The advantage of stepper motors that the position of the motor is "known". Zero position can be determined, if the original position is known.

Stepping motors come in a wide range of angular resolution and the coarsest motors typically turn 90 degrees per step. High resolution permanent magnet motors are only able to handle about 18 degrees less than that. With the right controller stepper motors can be run in half-steps, which is amazing.

The main complaint about the stepper motor is that it usually draws more power than a standard DC motor and maneuvering is also difficult.

About the Author:
Linear Motors and Stepper Motors

Article Source: http://www.articlesbase.com/technology-articles/linear-stepper-motors-technology-31692.html

Operating Analysis of Different Stepping Motor Control Mechanisms

Author: s.sankar

This section covers all types of motors, from the elementary circuitry needed to control a variable reluctance motor, to the H-bridge circuitry needed to control a bipolar permanent magnet motor. Each class of drive circuit is illustrated with practical examples, but these examples are not intended as an exhaustive catalog of the commercially available control circuits, nor is the information given here intended to substitute for the information found on the manufacturer's component data sheets for the parts mentioned.

This section only covers the most elementary control circuitry for each class of motor. All of these circuits assume that the motor power supply provides a drive voltage no greater than the motor's rated voltage, and this significantly limits motor performance. The next section, on current limited drive circuitry, covers practical high-performance drive circuits.

Variable Reluctance Motors

Typical controllers for variable reluctance stepping motors are variations on the outline shown in Figure 3.1:

Figure 3.1

In Figure 3.1, boxes are used to represent switches; a control unit, not shown, is responsible for providing the control signals to open and close the switches at the appropriate times in order to spin the motors. In many cases, the control unit will be a computer or programmable interface controller, with software directly generating the outputs needed to control the switches, but in other cases, additional control circuitry is introduced, sometimes gratuitously!

Motor windings, solenoids and similar devices are all inductive loads. As such, the current through the motor winding cannot be turned on or off instantaneously without involving infinite voltages! When the switch controlling a motor winding is closed, allowing current to flow, the result of this is a slow rise in current. When the switch controlling a motor winding is opened, the result of this is a voltage spike that can seriously damage the switch unless care is taken to deal with it appropriately.

There are two basic ways of dealing with this voltage spike. One is to bridge the motor winding with a diode, and the other is to bridge the motor winding with a capacitor. Figure 3.2 illustrates both approaches:

Figure 3.2

The diode shown in Figure 3.2 must be able to conduct the full current through the motor winding, but it will only conduct briefly each time the switch is turned off, as the current through the winding decays. If relatively slow diodes such as the common 1N400X family are used together with a fast switch, it may be necessary to add a small capacitor in parallel with the diode.

The capacitor shown in Figure 3.2 poses more complex design problems! When the switch is closed, the capacitor will discharge through the switch to ground, and the switch must be able to handle this brief spike of discharge current. A resistor in series with the capacitor or in series with the power supply will limit this current. When the switch is opened, the stored energy in the motor winding will charge the capacitor up to a voltage significantly above the supply voltage, and the switch must be able to tolerate this voltage. To solve for the size of the capacitor, we equate the two formulas for the stored energy in a resonant circuit:

P = C V2 / 2

P = L I2 / 2

Where:

P -- stored energy, in watt seconds or coulomb volts

C -- capacity, in farads

V -- voltage across capacitor

L -- inductance of motor winding, in henrys

I -- current through motor winding

Solving for the minimum size of capacitor required to prevent overvoltage on the switch is fairly easy:

C > L I2 / (Vb - Vs)2

Where:

Vb -- the breakdown voltage of the switch

Vs -- the supply voltage

Variable reluctance motors have variable inductance that depends on the shaft angle. Therefore, worst-case design must be used to select the capacitor. Furthermore, motor inductances are frequently poorly documented, if at all.

The capacitor and motor winding, in combination, form a resonant circuit. If the control system drives the motor at frequencies near the resonant frequency of this circuit, the motor current through the motor windings, and therefore, the torque exerted by the motor, will be quite different from the steady state torque at the nominal operating voltage! The resonant frequency is:

f = 1 / ( 2 (L C)0.5 )

Again, the electrical resonant frequency for a variable reluctance motor will depend on shaft angle! When a variable reluctance motors is operated with the exciting pulses near resonance, the oscillating current in the motor winding will lead to a magnetic field that goes to zero at twice the resonant frequency, and this can severely reduce the available torque!

Unipolar Permanent Magnet and Hybrid Motors

Typical controllers for unipolar stepping motors are variations on the outline shown in Figure 3.3:

Figure 3.3

In Figure 3.3, as in Figure 3.1, boxes are used to represent switches; a control unit, not shown, is responsible for providing the control signals to open and close the switches at the appropriate times in order to spin the motors. The control unit is commonly a computer or programmable interface controller, with software directly generating the outputs needed to control the switches.

As with drive circuitry for variable reluctance motors, we must deal with the inductive kick produced when each of these switches is turned off. Again, we may shunt the inductive kick using diodes, but now, 4 diodes are required, as shown in Figure 3.4:

Figure 3.4

The extra diodes are required because the motor winding is not two independent inductors, it is a single center-tapped inductor with the center tap at a fixed voltage. This acts as an autotransformer! When one end of the motor winding is pulled down, the other end will fly up, and visa versa. When a switch opens, the inductive kickback will drive that end of the motor winding to the positive supply, where it is clamped by the diode. The opposite end will fly downward, and if it was not floating at the supply voltage at the time, it will fall below ground, reversing the voltage across the switch at that end. Some switches are immune to such reversals, but others can be seriously damaged.

A capacitor may also be used to limit the kickback voltage, as shown in Figure 3.5:

Figure 3.5

The rules for sizing the capacitor shown in Figure 3.5 are the same as the rules for sizing the capacitor shown in Figure 3.2, but the effect of resonance is quite different! With a permanent magnet motor, if the capacitor is driven at or near the resonant frequency, the torque will increase to as much as twice the low-speed torque! The resulting torque versus speed curve may be quite complex, as illustrated in Figure 3.6:

Figure 3.6

Figure 3.6 shows a peak in the available torque at the electrical resonant frequency, and a valley at the mechanical resonant frequency. If the electrical resonant frequency is placed appropriately above what would have been the cutoff speed for the motor using a diode-based driver, the effect can be a considerable increase in the effective cutoff speed.

The mechanical resonant frequency depends on the torque, so if the mechanical resonant frequency is anywhere near the electrical resonance, it will be shifted by the electrical resonance! Furthermore, the width of the mechanical resonance depends on the local slope of the torque versus speed curve; if the torque drops with speed, the mechanical resonance will be sharper, while if the torque climbs with speed, it will be broader or even split into multiple resonant frequencies.

Practical Unipolar and Variable Reluctance Drivers

In the above circuits, the details of the necessary switches have been deliberately ignored. Any switching technology, from toggle switches to power MOSFETS will work! Figure 3.7 contains some suggestions for implementing each switch, with a motor winding and protection diode included for orientation purposes:

Figure 3.7

Each of the switches shown in Figure 3.7 is compatible with a TTL input. The 5 volt supply used for the logic, including the 7407 open-collector driver used in the figure, should be well regulated. The motor power, typically between 5 and 24 volts, needs only minimal regulation. It is worth noting that these power switching circuits are appropriate for driving solenoids, DC motors and other inductive loads as well as for driving stepping motors.

The SK3180 transistor shown in Figure 3.7 is a power darlington with a current gain over 1000; thus, the 10 milliamps flowing through the 470 ohm bias resistor is more than enough to allow the transistor to switch a few amps current through the motor winding. The 7407 buffer used to drive the darlington may be replaced with any high-voltage open collector chip that can sink at least 10 milliamps. In the event that the transistor fails, the high-voltage open collector driver serves to protects the rest of the logic circuitry from the motor power supply.

The IRC IRL540 shown in Figure 3.7 is a power field effect transistor. This can handle currents of up to about 20 amps, and it breaks down nondestructively at 100 volts; as a result, this chip can absorb inductive spikes without protection diodes if it is attached to a large enough heat sink. This transistor has a very fast switching time, so the protection diodes must be comparably fast or bypassed by small capacitors. This is particularly essential with the diodes used to protect the transistor against reverse bias! In the event that the transistor fails, the zener diode and 100 ohm resistor protect the TTL circuitry. The 100 ohm resistor also acts to somewhat slow the switching times on the transistor.

For applications where each motor winding draws under 500 milliamps, the ULN200x family of darlington arrays from Allegro Microsystems, also available as the DS200x from National Semiconductor and as the Motorola MC1413 darlington array will drive multiple motor windings or other inductive loads directly from logic inputs. Figure 3.8 shows the pinout of the widely available ULN2003 chip, an array of 7 darlington transistors with TTL compatible inputs:

Figure 3.8

The base resistor on each darlington transistor is matched to standard bipolar TTL outputs. Each NPN darlington is wired with its emitter connected to pin 8, intended as a ground pin, Each transistor in this package is protected by two diodes, one shorting the emitter to the collector, protecting against reverse voltages across the transistor, and one connecting the collector to pin 9; if pin 9 is wired to the positive motor supply, this diode will protect the transistor against inductive spikes.

The ULN2803 chip is essentially the same as the ULN2003 chip described above, except that it is in an 18-pin package, and contains 8 darlingtons, allowing one chip to be used to drive a pair of common unipolar permanent-magnet or variable-reluctance motors.

For motors drawing under 600 milliamps per winding, the UDN2547B quad power driver made by Allegro Microsystems will handle all 4 windings of common unipolar stepping motors. For motors drawing under 300 milliamps per winding, Texas Instruments SN7541, 7542 and 7543 dual power drivers are a good choice; both of these alternatives include some logic with the power drivers.

Bipolar Motors and H-Bridges

Things are more complex for bipolar permanent magnet stepping motors because these have no center taps on their windings. Therefore, to reverse the direction of the field produced by a motor winding, we need to reverse the current through the winding. We could use a double-pole double throw switch to do this electromechanically; the electronic equivalent of such a switch is called an H-bridge and is outlined in

Figure 3.9

As with the unipolar drive circuits discussed previously, the switches used in the H-bridge must be protected from the voltage spikes caused by turning the power off in a motor winding. This is usually done with diodes, as shown in Figure 3.9.

It is worth noting that H-bridges are applicable not only to the control of bipolar stepping motors, but also to the control of DC motors, push-pull solenoids (those with permanent magnet plungers) and many other applications.

With 4 switches, the basic H-bridge offers 16 possible operating modes, 7 of which short out the power supply! The following operating modes are of interest:

Forward mode, switches A and D closed.

Reverse mode, switches B and C closed.

These are the usual operating modes, allowing current to flow from the supply, through the motor winding and onward to ground. Figure 3.10 illustrates forward mode:

Figure 3.10

Fast decay mode or coasting mode, all switches open.

Any current flowing through the motor winding will be working against the full supply voltage, plus two diode drops, so current will decay quickly. This mode provides little or no dynamic braking effect on the motor rotor, so the rotor will coast freely if all motor windings are powered in this mode. Figure 3.11 illustrates the current flow immediately after switching from forward running mode to fast decay mode.

Figure 3.11

Slow decay modes or dynamic braking modes.

In these modes, current may recirculate through the motor winding with minimum resistance. As a result, if current is flowing in a motor winding when one of these modes is entered, the current will decay slowly, and if the motor rotor is turning, it will induce a current that will act as a brake on the rotor. Figure 3.12 illustrates one of the many useful slow-decay modes, with switch D closed; if the motor winding has recently been in forward running mode, the state of switch B may be either open or closed:

Figure 3.12

Most H-bridges are designed so that the logic necessary to prevent a short circuit is included at a very low level in the design. Figure 3.13 illustrates what is probably the best arrangement:

Figure 3.13

Here, the following operating modes are available:

XY ABCD Mode

00 0000 fast decay

01 1001 forward

10 0110 reverse

11 0101 slow decay

The advantage of this arrangement is that all of the useful operating modes are preserved, and they are encoded with a minimum number of bits; the latter is important when using a microcontroller or computer system to drive the H-bridge because many such systems have only limited numbers of bits available for parallel output. Sadly, few of the integrated H-bridge chips on the market have such a simple control scheme.

Practical Bipolar Drive Circuits

There are a number of integrated H-bridge drivers on the market, but it is still useful to look at discrete component implementations for an understanding of how an H-bridge works. Antonio Raposo (ajr@cybill.inesc.pt) suggested the H-bridge circuit shown in Figure 3.14;

Figure 3.14

The X and Y inputs to this circuit can be driven by open collector TTL outputs as in the darlington-based unipolar drive circuit in Figure 3.7. The motor winding will be energised if exactly one of the X and Y inputs is high and exactly one of them is low. If both are low, both pull-down transistors will be off. If both are high, both pull-up transistors will be off. As a result, this simple circuit puts the motor in dynamic braking mode in both the 11 and 00 states, and does not offer a coasting mode.

The circuit in Figure 3.14 consists of two identical halves, each of which may be properly described as a push-pull driver. The term half H-bridge is sometimes applied to these circuits! It is also worth noting that a half H-bridge has a circuit quite similar to the output drive circuit used in TTL logic. In fact, TTL tri-state line drivers such as the 74LS125A and the 74LS244 can be used as half H-bridges for small loads, as illustrated in Figure 3.15:

Figure 3.15

This circuit is effective for driving motors with up to about 50 ohms per winding at voltages up to about 4.5 volts using a 5 volt supply. Each tri-state buffer in the LS244 can sink about twice the current it can source, and the internal resistance of the buffers is sufficient, when sourcing current, to evenly divide the current between the drivers that are run in parallel. This motor drive allows for all of the useful states achieved by the driver in Figure 3.13, but these states are not encoded as efficiently:

XYE Mode

--1 fast decay

000 slower decay

010 forward

100 reverse

110 slow decay

The second dynamic braking mode, XYE=110, provides a slightly weaker braking effect than the first because of the fact that the LS244 drivers can sink more current than they can source.

The Microchip (formerly Telcom Semiconductor) TC4467 Quad CMOS driver is another example of a general purpose driver that can be used as 4 independent half H-bridges. Unlike earlier drivers, the data sheet for this driver even suggests using it for motor control applications, with supply voltages up to 18 volts and up to 250 milliamps per motor winding.

One of the problems with commercially available stepping motor control chips is that many of them have relatively short market lifetimes. For example, the Seagate IPxMxx series of dual H-bridge chips (IP1M10 through IP3M12) were very well thought out, but unfortunately, it appears that Seagate only made these when they used stepping motors for head positioning in Seagate disk drives. The Toshiba TA7279 dual H-bridge driver would be another another excellent choice for motors under 1 amp, but again, it appears to have been made for internal use only.

The SGS-Thompson (and others) L293 dual H-bridge is a close competitor for the above chips, but unlike them, it does not include protection diodes. The L293D chip, introduced later, is pin compatible and includes these diodes. If the earlier L293 is used, each motor winding must be set across a bridge rectifier (1N4001 equivalent). The use of external diodes allows a series resistor to be put in the current recirculation path to speed the decay of the current in a motor winding when it is turned off; this may be desirable in some applications. The L293 family offers excellent choices for driving small bipolar steppers drawing up to one amp per motor winding at up to 36 volts. Figure 3.16 shows the pinout common to the L293B and L293D chips:

Figure 3.16

This chip may be viewed as 4 independent half H-bridges, enabled in pairs, or as two full H-bridges. This is a power DIP package, with pins 4, 5, 12 and 13 designed to conduct heat to the PC board or to an external heat sink.

The SGS-Thompson (and others) L298 dual H-bridge is quite similar to the above, but is able to handle up to 2-amps per channel and is packaged as a power component; as with the LS244, it is safe to wire the two H-bridges in the L298 package into one 4-amp H-bridge (the data sheet for this chip provides specific advice on how to do this). One warning is appropriate concerning the L298; this chip very fast switches, fast enough that commonplace protection diodes (1N400X equivalent) don't work. Instead, use a diode such as the BYV27. The National Semiconductor LMD18200 H-bridge is another good example; this handles up to 3 amps and has integral protection diodes.

While integrated H-bridges are not available for very high currents or very high voltages, there are well designed components on the market to simplify the construction of H-bridges from discrete switches. For example, International Rectifier sells a line of half H-bridge drivers; two of these chips plus 4 MOSFET switching transistors suffice to build an H-bridge. The IR2101, IR2102 and IR2103 are basic half H-bridge drivers. Each of these chips has 2 logic inputs to directly control the two switching transistors on one leg of an H-bridge. The IR2104 and IR2111 have similar output-side logic for controlling the switches of an H-bridge, but they also include input-side logic that, in some applications, may reduce the need for external logic. In particular, the 2104 includes an enable input, so that 4 2104 chips plus 8 switching transistors can replace an L293 with no need for additional logic.

The data sheet for the Microchip (formerly Telcom Semiconductor) TC4467 family of quad CMOS drivers includes information on how to use drivers in this family to drive the power MOSFETs of H-bridges running at up to 15 volts.

A number of manufacturers make complex H-bridge chips that include current limiting circuitry; these are the subject of the next section. It is also worth noting that there are a number of 3-phase bridge drivers on the market, appropriate for driving Y or delta configured 3-phase permanent magnet steppers. Few such motors are available, and these chips were not developed with steppers in mind. Nonetheless, the Toshiba TA7288P, the GL7438, the TA8400 and TA8405 are clean designs, and 2 such chips, with one of the 6 half-bridges ignored, will cleanly control a 5-winding 10 step per revolution motor.

About the Author:

Assistant professor in lord venkateswara engineering college.I am doing phd in sathyabama university, Tamil Nadu,India.

Article Source: http://www.articlesbase.com/electronics-articles/operating-analysis-of-different-stepping-motor-control-mechanisms-590072.html

Linear Motors and Stepper Motors ,Open Loop Solutions and Current Limiting for Stepping Motors

Linear Motors and Stepper Motors
By Gordon Petten

A linear induction motor is made up of an inductor which is made of individual cores with a concentrated polyphase. Linear motors can be directly substituted for ball screw drives, hydraulic drives, pneumatic drives, or cam drives.

A linear induction motor is basically what is referred to by experts as a “rotating squirrel cage” induction motor. The difference is that the motor is opened out flat. Instead of producing rotary torque from a cylindrical machine it produces linear force from a flat machine. The shape and the way it produces motion is changed, however it is still the same as its cylindrical counterpart. There are no moving parts, however and most experts don’t like that. It does have a silent operation and reduced maintenance as well as a compact size, which appeals many engineers. There is also a universal agreement that it has an ease of control and installation. These are all important considerations when thinking about what type of device you want to create. The linear induction motor thrusts ratio varies depending mainly on the size and rating. Speeds of the linear induction motor vary from zero to many meters per second. Speed can be controlled. Stopping, starting and reversing are all easy. Linear induction motors are improving constantly and with improved control, lower life cycle cost, reduced maintenance and higher performance they are becoming the choice of the experts. Linear motors are simple to control and easy to use. They have a fast response and high acceleration. Their speed is not dependant on contact friction so it is easier to pick up speed quickly.

Stepper motors are a special kind of motor that moves in discrete steps. When one set of windings is energized the motor moves a step in one direction and when another set of windings is energized the motor moves a step in the other direction. The advantage of stepper motors that the position of the motor is "known". Zero position can be determined, if the original position is known.

Stepping motors come in a wide range of angular resolution and the coarsest motors typically turn 90 degrees per step. High resolution permanent magnet motors are only able to handle about 18 degrees less than that. With the right controller stepper motors can be run in half-steps, which is amazing.

The main complaint about the stepper motor is that it usually draws more power than a standard DC motor and maneuvering is also difficult. Rotary Tables

Linear Actuators

Article Source: http://EzineArticles.com/?expert=Gordon_Petten

Open Loop Solutions and Current Limiting for Stepping Motors

Author: s.sankar

There is good reason to run a stepping motor at a supply voltage above that needed to push the maximum rated current through the motor windings. Running a motor at higher voltages leads to a faster rise in the current through the windings when they are turned on, and this, in turn, leads to a higher cutoff speed for the motor and higher torques at speeds above the cutoff.

Microstepping, where the control system positions the motor rotor between half steps, also requires external current limiting circuitry. For example, to position the rotor 1/4 of the way from one step to another, it might be necessary to run one motor winding at full current while the other is run at approximately 1/3 of that current.

The remainder of this section discusses various circuits for limiting the current through the windings of a stepping motor, starting with simple resistive limiters and moving up to choppers and other switching regulators. Most of these current limiters are appropriate for many other applications, including limiting the current through conventional DC motors and other inductive loads.

Resistive Current Limiters

The easiest to understand current limiter is a series resistor. Most motor manufacturers recommended this approach in their literature up until the early 1980's, and most motor data sheets still give performance curves for motors driven by such circuits. The typical circuits used to control the current through one winding of a permanent magnet or hybrid motor are shown in Figure 4.1.

Figure 4.1

R1 in this figure limits the current through the motor winding. Given a rated current of I and a motor winding with a resistance Rw, Ohm's law sets the maximum supply voltage as I(Rw+R1). Given that the inductance of the motor motor winding is Lw, the time constant for the motor winding will be Lw/(Rw+R1). Figure 4.2 illustrates the effect of increasing the resistance and the operating voltage on the rise and fall times of the current through one winding of a stepping motor.

Figure 4.2

R2 is shown only in the unipolar example in Figure 4.1 because it is particularly useful there. For a bipolar H-bridge drive, when all switches are turned off, current flows from ground to the motor supply through R1, so the current through the motor winding will decay quite quickly. In the unipolar case, R2 is necessary to equal this performance. When the switches in the H-bridge circuit shown in Figure 4.1 are opened, the direction of current flow through R1 will reverse almost instantaneously! If R1 has any inductance, for example, if it is wire-wound, it must either be bypassed with a capacitor to handle the voltage kick caused by this current reversal, or R2 must be added to the H-bridge.

Given the rated maximum current through each winding and the supply voltage, the resistance and wattage of R1 is easy to compute. R2 if it is included, poses more interesting problems. The resistance of R2 depends on the maximum voltage the switches can handle. For example, if the supply voltage is 24 volts, and the switches are rated at 75 volts, the drop across R2 can be as much as 51 volts without harming the transistors. Given an operating current of 1.5 amps, R2 can be a 34 ohm resistor. Note that an interesting alternative is to use a zener diode in place of R2.

Figuring the peak average power R2 must dissipate is a wonderful exercise in dynamics; the inductance of the motor windings is frequently undocumented and may vary with the rotor position. The power dissipated in R2 also depends on the control system. The worst case occurs when the control system chops the power to one winding at a high enough frequency that the current through the motor winding is effectively constant; the maximum power is then a function of the duty cycle of the chopper and the ratios of the resistances in the circuit during the on and off phases of the chopper. Under normal operating conditions, the peak power dissipation will be significantly lower.

Linear Current Limiters

A pair of high wattage power resistors can cost more than a pair of power transistors plus a heat sink, particularly if forced air cooling is available. Furthermore, a transistorized constant current source, as shown in Figure 4.3, will give faster rise times through the motor windings than the current limiting resistor shown in Figure 4.1. This is because a current source will deliver the full supply voltage across the motor winding until the current reaches the rated current; only then will the current source drop the voltage.

Figure 4.3

In Figure 4.3, a transistorized current source (T1 plus R1) has been substituted for the current limiting resistor R1 used in the examples in Figure 4.1. The regulated voltage supplied to the base of T1 serves to regulate the voltage across the sense resistor R1, and this, in turn, maintains a constant current through R1 so long as any current is allowed to flow through the motor winding. Typically, R1 will have as low a resistance as possible, in order to avoid the high cost of a power resistor. For example, if the forward voltage drops across the diode in series with the base T1 and VBE for T1 are both 0.65 volts, and if a 3.3 volt zener diode is used for a reference, the voltage across R1 will be maintained at about 2.0 volts, so if R1 is 2 ohms, this circuit will limit the current to 1 amp, and R1 must be able to handle 2 watts. R3 in Figure 4.3 must be sized in terms of the current gain of T1 so that sufficient current flows through R1 and R3 to allow T1 to conduct the full rated motor current.

The transistor T1 used as a current regulator in Figure 4.3 is run in linear mode, and therfore, it must dissipate quite a bit of power. For example, if the motor windings have a resistance of 5 ohms and a rated current of 1 amp, and a 25 volt power supply is used, T1 plus R1 will dissapate, between them, 20 watts! The circuits discussed in the following sections avoid this waste of power while retaining the performance advantages of the circuit given here.

When an H-bridge bipolar drive is used with a resistive current limiter, as shown in Figure 4.1, the resistor R2 was not needed because current could flow backwards through R1. When a transistorized current limiter is used, current cannot flow backwards through T1, so a separate current path back to the positive supply must be provided to handle the decaying current through the motor windings when the switches are opened. R2 serves this purpose here, but a zener diode may be substituted to provide even faster turn-off.

The performance of a motor run with a current limited power supply is noticably better than the performance of the same motor run with a resistively limited supply, as illustrated in Figure 4.4:

Figure 4.4

With either a current limited supply or a resistive current limiter, the initial rate of increase of the current through the inductive motor winding when the power is turned on depends only on the inductance of the winding and the supply voltage. As the current increases, the voltage drop across a resistive current limiter will increase, dropping the voltage applied to the motor winding, and therefore, dropping the rate of increase of the current through the winding. As a result, the current will only approach the rated current of the motor winding asymptotically In contrast, with a pure current limiter, the current through the motor winding will increase almost linearly until the current limiter cuts in, allowing the current to reach the limit value quite quickly. In fact, the current rise is not linear; rather, the current rises asymptotically towards a limit established by the resistance of the motor winding and the resistance of the sense resistor in the current limiter. This maximum is usually well above the rated current for the motor winding.

Open Loop Current Limiters

Both the resistive and the linear transistorized current limiters discussed above automatically limit the current through the motor winding, but at a considerable cost, in terms of wasted heat. There are two schemes that eliminate this expense, although at some risk because of the lack of feeback about the current through the motor.

Use of a Voltage Boost

If you plot the voltage across the motor winding as a function of time, assuming the use of a transistorized current limiter such as is illustrated in Figure 4.3, and assuming a 1 amp 5 ohm motor winding, the result will be something like that illustrated in Figure 4.5:

Figure 4.5

As long as the current is below the current limiter's set point, almost the full supply voltage is applied across the motor winding. Once the current reaches the set point, the voltage across the motor winding falls to that needed to sustain the current at the set point, and when the switches open, the voltage reverses briefly as current flows through the diode network and R2. An alternative way to get this voltage profile is to use a dual-voltage power supply, turning on the high voltage for as long as it takes to bring the current in the motor winding up to the rated current, and then turning off the high voltage and turning on the sustaining voltage. Some motor controllers do this directly, without monitoring the current through the motor windings. This provides excellent performance and minimizes power losses in the regulator, but it offers a dangerous temptation.

If the motor does not deliver enough torque, it is tempting to simply lengthen the high-voltage pulse at the time the motor winding is turned on. This will usually provide more torque, although saturation of the magnetic circuits frequently leads to less torque than might be expected, but the cost is high! The risk of burning out the motor is quite real, as is the risk of demagnitizing the motor rotor if it is turned against the imposed field while running hot. Therefore, if a dual-voltage supply is used, the temptation to raise the torque in this way should be avoided!

The problems with dual voltage supplies are particularly serious when the time intervals are under software control, because in this case, it is common for the software to be written by a programmer who is insufficiently aware of the physical and electrical characteristics of the control system.

Use of Pulse Width Modulation

Another alternative approach to controlling the current through the motor winding is to use a simple power supply controlled by pulse width modulaton (PWM) or by a chopper. During the time the current through the motor winding is increasing, the control system leaves the supply attached with a 100% duty cycle. Once the current is up to the full rated current, the control system changes the duty cycle to that required to maintain the current. Figure 4.6 illustrates this scheme:

Figure 4.6

For any chopper or pulse width modulator, we can define the duty-cycle D as the fraction of each cycle that the switch is closed:

D = Ton / (Ton + Toff)

Where

Ton -- time the switch is closed during each cycle

Toff -- time the switch is open during each cycle

The voltage curve shown above indicates the full supply voltage being applied to the motor winding during the on-phase of every chopper cycle, while when the chopper is off, a negative voltage is shown. This is the result of the forward voltage drop in the diodes that are used to shunt the current when the switches turn off, plus the external resistance used to speed the decay of the current through the motor winding. For large values of Ton or Toff, the exponential nature of the rise and fall of the current through the motor winding is significant, but for sufficiently small values, we can approximate these as linear. Assuming that the chopper is working to maintain a current of I and that the amplitude is small, we will approximate the rates of rise and fall in the current in terms of the voltage across the motor winding when the switch is closed and when it is open:

Von = Vsupply - I(Rwinding + Ron)

Voff = Vdiode + I(Rwinding + Roff)

Here, we lump together all resistances in series with the winding and power supply in the on state as Ron, and we lump together all resistances in the current recirculation path when the switch(es) are open as Roff. The forward voltage drops of any diodes in the current recirculation path have been lumped as Vdiode; if the off-state recirculation path runs from ground to the power supply (H-bridge fast decay mode), the supply voltage must also be included in Vdiode. Forward voltage drops of any switches in the on-state and off-state paths should also be incorporated into these voltages.

To solve for the duty cycle, we first note that:

dI/dt = V/L

Where

I -- current through the motor winding

V -- voltage across the winding

L -- inductance of the winding

We then substitute the specific voltages for each phase of operation:

Iripple / Toff = Voff / L

Iripple / Ton = Von / L

Where

Iripple -- the peak to peak ripple in the current

Solving for Toff and Ton and then substituting these into the definition of the duty cycle of the chopper, we get:

D = Ton / (Ton + Toff) = Voff / (Von + Voff)

If the forward voltage drops in diodes and switches are negligable, and if the only significant resistance is that of the motor winding itself, this simplifies to:

D = I Rwinding / Vsupply = Vrunning / Vsupply

This special case is particularly desirable because it delivers all of the power to the motor winding, with no losses in the regulation system, without regard for the difference between the supply voltage and the running voltage.

The AC ripple Iripple superimposed on the running current by a chopper can be a source of problems; at high frequencies, it can be a source of RF emissions, and at audio frequencies, it can be a source of annoying noise. For example, with audio frequency chopping, most stepper controlled systems will "squeel", sometimes loudly, when the rotor is displaced from the equilibrium position. For small systems, this is usually no more than a minor nuisance, but in systems with large numbers of high power steppers, the ripple currents can induce dangerous AC voltages on nearby signal lines and dangerous currents in nearby ground lines. To find the ripple amplitude, first recall that:

Iripple / Toff = Voff / L

Then solve for Iripple:

Iripple = Toff Voff / L

Thus, to reduce the ripple amplitude at any particular duty cycle, it is necessary to increase the chopper frequency. This cannot be done without limit because switching losses increase with frequency. Note that this change has no significant effect on AC losses; the decrease in such losses due to decreased amplitude in the ripple is generally offset by the effect of increasing frequency.

The primary problem with use of a simple chopping or pulse-width modulation control scheme is that it is completely open loop. Design of good chopper based control systems requires knowledge of motor characteristics such as inductance that are frequently poorly documented, and as with dual-voltage supplies, when motor performance is marginal, it is very tempting to increase the duty-cycle without attention to the long-term effects of this on the motor. In the designs that follow, this weakness will be addressed by introducing feedback loops into the low level drive system to directly monitor the current and determine the duty cycle.

One-Shot Feedback Current Limiting

The most common approach to automatically adjusting the duty cycle of the switches in the stepper driver involves monitoring the current to the motor windings; when it rises too high, the winding is turned off for a fixed interval. This requires a current sensing system and a one-shot, as illustrated in Figure 4.7:

Figure 4.7

Figure 4.7 illustrates a unipolar drive system. As with the circuit given in Figure 4.3, R1 should be as small as possible, limited only by the requirement that the sense voltage provided to the comparator must be high enough to be within its operating range. Note that when the one-shot output (¬Q) is low, the voltage across R1 no-longer reflects the current through the motor winding. Therefore, the one-shot must be insensitive to the output of the comparator between the time it fires and the time it resets. Practical circuit designs using this approach involve some complexity to meet this constraint! Selecting the value of R2 for the circuit shown in Figure 4.7 poses problems. If R2 is large, the current through the motor windings will decay quickly when the higher level control system turns off this motor winding, but when the winding is turned on, the current ripple will be large and the power lost in R2 will be significant. If R2 is small, this circuit will be very energy efficient but the current through the motor winding will decay only slowly when this winding is turned off, and this will reduce the cutoff speed for the motor.

The peak power dissipated in R2 will be I2R2 during Toff and zero during Ton; thus, the average power dissipated in R2 when the motor winding is on will be:

P2 = I2R Toff / (Ton + Toff)

Recall that the duty cycle D is defined as Ton/(Ton+Toff) and may be approximated as Vrunning/Vsupply. As a result, we can approximate the power dissipation as:

P2 = I2R2 (1 - Vrunning/Vsupply).

Given the usual safety margins used in selecting power resistor wattages, a better approximation is not necessary.

When designing a control system based on pulse width modulation, note that the cutoff time for the one-shot determines Toff, and that this is fixed, determined by the timing network attached to the one-shot. Ideally, this should be set as follows:

Toff = L Iripple / Voff

This presumes that the inductance L of the motor winding is known, that the acceptable magnitude of Iripple is known, and that Voff, the total reverse voltage in the current recirculation path, is known and fixed. Note that this scheme leads to a variable chopping rate. As with the linear current limiters shown in Figure 4.3, the full supply voltage will be applied during the turn-on phase, and the chopping action only begins when the motor winding reaches the current limit set by Vref. This circuit will vary the chopping rate to compensate for changes in the back EMF of the motor winding, for example, those caused by rotor motion; in this regard, it offers the same quality of regulation as the linear current limiter. The one-shot current regulator shown in Figure 4.7 can also be applied to an H-bridge regulator. The encoded H-bridge shown in Figure 3.13 is an excellent candidate for this application, as shown in Figure 4.8:

Figure 4.8

Unlike the circuit in Figure 4.7, this circuit does not provide design tradeoffs in the selection of the resistance in the current decay path; instead, it offers the same selection of decay paths as was available in the original circuit from Figure 3.13. If the X and Y control inputs are held in a running mode (01 or 10), the current limiter will alternate between that running and slow decay modes, maximizing energy efficiency. When the time comes to turn off the current through the motor winding, the X and Y inputs may be set to 00, using fast decay mode to maximize the cutoff speed, while if the damping effect of dynamic braking is needed to control resonance, X and Y may be set to 11.

Note that the current recirculation path during dynamic braking does not pass through R1, and as a result, if the motor generates a large amount of power, burnt out components in the motor or controller are likely. This is unlikely to cause problems with stepping motors, but when dynamic braking is used with DC motors, the current limiter should be arranged to remain engaged while in braking mode!

Practical Examples

SGS-Thompson (and others) L293 (1A) and L298 (2A) dual H-bridges are designed for easy use with partial feedback current limiters. These chips have enable inputs for each H-bridge that can be directly connected to the output of the one-shot, and they have ground connections for motor-power that are isolated from their logic ground connections; this allows sense resistors to be easily incorporated into the circuit. The 3952 H-bridge from Allegro Microsystems can handle up to 2-amps at 50 volts and incorporates all of the logic necessary for current control, including comparators and one-shot. This chip is available in many package styles; Figure 4.9 illustrates the DIP configuration wired for a constant current limit:

Figure 4.9

If Rt is 20 Kohm, and Ct is 1000pF, Toff for the pulse-width modulation will be fixed at 20 (±2) microseconds. The 3952 chip incorporates a 10 to 1 voltage divider on the Vref input, so attaching Vref to the 5 volt logic supply sets the actual reference voltage to 0.5 V. Thus, if the sense resistor Rs is 0.5 ohms, this arrangement will attempt to maintain a regulated current through the load of 1 A.

Note that all power switching chips are potentially serious sources of electromagnetic interfence! The 47µF capacitor shown between the motor power and ground should be as close to the chip as possible, and the path from the SENSE pin through Rs to ground and back to a ground pin of the chip should be very short and with a very low resistance.

On the 5 volt side, because Vref is taken from Vcc, a small decoupling capacitor should be placed very close to the chip. It may even be appropriate to isolate the Vref input from Vcc with a small series resistor and a separate decoupling capacitor. If this is done, note that the resistance from the Vref pin to ground through the chip's internal voltage divider is around 50 Kohms.

One of the more dismaying features of the 3952 chip, as well as many of its competitors, is the large number of control inputs. These are summarized in the following table:

BRAKE ENABLE PHASE MODE OUTa OUTb Notes

0 - - 0 0 0 Brake

0 - - 1 0 0 Limited Brake

1 1 - 0 - - Standby

1 1 - 1 - - Sleep

1 0 0 0 0 1 Reverse, Slow

1 0 0 1 0 1 Reverse, Fast

1 0 1 0 1 0 Forward, Slow

1 0 1 1 1 0 Forward, Fast

In the forward and reverse running modes, the mode input determines whether fast or slow decay modes are used during Toff. In the dynamic braking modes, the mode input determines whether the current limiter is enabled. This is of limited value with stepping motors, but use of dynamic braking without a current limiter can be dangerous with DC motors. In sleep mode, the power consumption of the chip is minimized. From the perspective of the load, sleep and standby modes put the load into fast decay mode (all switches off) but in sleep mode, the chip draws considerably less power, both from the logic supply and the motor supply.

Hysteresis Feedback Current Limiting

In many cases, motor control systems are expected to operate acceptably with a number of different stepping motors. The one-shot based current regulators illustrated in Figures 4.7 to 4.9 have an accuracy that depends on the inductance of the motor windings. Therefore, if fixed accuracy is required, any motor substation must be balanced by changes to the RC network that determines the off-time of the one-shot.

This section deals with alternative designs that eliminate the need for this tuning. These alternative designs offer fixed precision current regulation over a wide range of load inductances. The key to this approach is arrange the recirculation paths so that the current-sense resistor R1 is always in the circuit, and then turn the switches on or off depending only on the current.

The usually way to build this type of controller is to use a comparator with a degree of hysteresis, for example, by feeding the output of the comparator back into one of its inputs through a resistor network, as illustrated in Figure 4.10:

Figure 4.10

To compute the desired values of R2 and R3, we note that:

Vripple > Vhysteresis

Where:

Vripple = Iripple R1

Iripple -- the maximum ripple allowed in the current

and:

Vhysteresis = Vswing R2 / (R2 + R3)

Vswing -- the voltage swing at the output of the comparator

We can solve this for the ratio of the resistances:

R2 / (R2 + R3) <>

For example, if R1 is 0.5 ohms and we wish to regulate the current to within 10 milliamps, using a comparator with TTL compatable outputs and a voltage swing of 4 volts, the ratio must be no greater than .00125.

Note that the sum R2 + R3 determines the loading on Vref, assuming that the input resistance of the comparator is effectively infinite. Typically, therefore, this sum is made quite large.

One problem with the circuit given in Figure 4.10 is that it does not limit the current through the motor in dynamic braking or slow decay modes. Even if the current through the sense resistor vastly exceeds the desired current, switches B and D will remain closed in dynamic braking mode, and if the reference voltage is variable, rapid drops in the reference voltage will not be enforced by this control system.

The designers of the Allegro 3952 chip faced this problem, and passed the solution back to the user, providing a MODE input to determine whether the chopper alternated between running and fast decay mode or running and slow decay mode. Note that this chip uses a fixed off-time set by a one-shot, and therefore, switching between the two decay modes will change the precision of the current regulator. Given that such a change in precision is acceptable, we can modify the circuit from Figure 4.10 to automatically thrown the system into fast-decay mode if the running or dynamic braking current exceeds the set-point of the comparator by too great a margin. Figure 4.11 illustrates how this can be done using a second comparator:

Figure 4.11

As shown in Figure 4.11, the lower comparator directly senses the voltage across R1, while the upper comparator senses a higher voltage, determined by a resistor network. This network should hold the negative inputs of the two comparators just far enough apart to guarantee that, as the voltage across R1 rises, the top comparator will always open the top switches before the bottom comparator opens the bottom switches, and as the voltage across R1 falls, the bottom comparator will always close the bottom switches before the top comparator closes the top switches.

As a result, this system has two basic steady-state running modes. If the motor winding is drawing power, one of the bottom switches will remain closed while the opposite switch on the top is used to chop the power to the motor winding, alternating the state of the system between running and slow-decay mode.

If the motor winding is generating power, the top switches will remain open and the bottom switches will do the chopping, alternating between fast-decay and slow-decay modes as needed to keep the current within limits. If the two comparators have accuracies on the order of a millivolt with hysteresis on the order of 5 millivolts, it is reasonable to use a 5 millivolt difference between the top and bottom comparator. If we use the 5 volt logic supply as the pull-up supply for the resistor network, and we assume a nominal operating threshold of around 0.5 volts, the resistor network should have a ratio of 1:900; for example, a 90k resistor from +5 and a 100 ohm resistor between the two comparator inputs.

Practical Examples

The basic idea described in this section is also applicable to unipolar stepping motor controllers, although in this context, it is somewhat easier to apply if the reference voltage is measured with respect to the unregulated motor power supply. Figure 4.12 illustrates a practical example, using the forward voltage drop across an ordinary silicon diode as the reference voltage.

Figure 4.12

The circuit shown in Figure 4.12 uses a 2.4K resistor to provide a bias current of 10ma to the reference diode. A small capacitor should be added across the reference diode if the motor power supply is minimally regulated.

The 0.6 ohm value used for the current sensing resistor sets the regulator to 1 amp, assuming that the reference voltage is 0.6 volts. The 1000 to 1 ratio on the feedback network around the comparator sets the allowed ripple in the regulated current to around 8 ma.

The comparator shown in Figure 4.12 can be powered from the minimally regulated motor power supply, but only if it is able to operate with the inputs very close to its positive supply voltage. Although I have not tried it, the Mitsubishi M5249L comparator appears to be ideally suited to this job; it can work from a positive supply of up to 40 volts, and the input voltages are allowed to slightly exceed the positive supply voltage! The output of this comparator is open collector, so the hysteresis network shown in the figure also acts as a pull-up network, providing a pull-up current of a few milliamps. The diode to +5 shown in the figure clamps the comparator output to the logic supply voltage, protecting the and gate inputs from overvoltage.

Other Current Sensing Technologies

The feedback loops of all of the current limiters given above use the voltage drop across a small resistor to measure the current. This is an excellent choice for small motors, but it poses difficulties for large high-current motors! There are other current sensing technologies appropriate for such settings, most notably those that deliver only a fraction of the motor current to the sensing resistor, and those that measure the current by sensing the magnetic field around the conductor.

National Semiconductor had incorporated a very clever current sensor into a number of their H-bridges. This delivers a current to the sense resistor that is proportional to the current through the motor winding, but far lower. For example, on the LMD18200 H-bridge, the sense resistor receives exactly 377 micro amps per ampere flowing through the motor winding.

The key to the current sensing technology used in the National Semiconductor line of H-bridges is found in the internal structure of the DMOS power switching transistors they use. These transistors are composed of thousands of small MOSFET transistor cells wired in parallel. A small but representative fraction of these cells, typically 1 in 4000, is used to extract the sense current while the remainder of the cells controls the motor current. The data sheet for the National LMD18245 H-bridge contains an excellent write-up on how this is done.

When very high currents are involved, precluding use of an integrated H-bridge, an appealing and well established current sensing technology involves the use of a split ferrite core and a hall effect sensor, as illustrated in Figure 4.13:

Figure 4.13

Simple linear Hall effect sensors require a small regulated bias current between two of their terminals, and they generate a DC voltage proportional to the magnetic field on a third terminal. The magnetic field across the gap sawed in the ferrite core is proportional to the current through the wire, and therefore, the voltage reported by the Hall effect sensor will be proportional to the current.

Allegro Microsystems and others make a full lines of Hall effect sensors, but pre-calibrated hall effect current sensors are available; these include the split core, the hall effect sensor, and auxiliary components, all mounted on a small PC board or potted as a unit. Newark Electronics lists a few sources of these, including Honeywell, F. W. Bell and LEM Instruments.

An intriguing new current sensor is just becoming available, as of 1998, based on a thin-film magneto resistive sensor; the sensitivity of this technology eliminates the need for the ferrite core and the result is a very compact current sensor. The NT series sensors made by F. W. Bell use this technology.

About the Author:

Assistant professor in lord venkateswara engineering college.I am doing phd in sathyabama university, Tamil Nadu,India.

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